Solid-state RF power amplifier for radio transmitters

ABSTRACT

An RF power amplifier includes a push-pull amplifier having field effect transistors. Temperature compensating bias circuitry provides a temperature compensated bias voltage to the transistors for decreasing the bias voltage thereof as temperature increases. The temperature compensating bias circuitry includes a temperature sensor generating a temperature signal. A first amplifier provides a first temperature dependent voltage based on the temperature signal. A second amplifier provides a second temperature dependent voltage based on the temperature signal. The first and second temperature dependent voltages change at substantially the same rate in response to the temperature signal. A potentiometer receives the first and second temperature dependent voltages such that a voltage across the potentiometer remains substantially constant when the first and second temperature dependent voltages change. An output of the bias circuitry is connected to at least one of the transistors and supplies the temperature compensated bias voltage to the at least one of the transistors.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a continuation-in-part of U.S. patent applicationSer. No. 11/749,786, filed on May 17, 2007, which claims the benefit ofU.S. Provisional Patent Application Ser. No. 60/801,006, filed May 17,2006 and the benefit of U.S. Provisional Patent Application Ser. No.60/747,662, filed May 18, 2006. The respective disclosures in theabove-referenced applications are incorporated herein by reference intheir entireties.

BACKGROUND OF THE INVENTION

1. Field of the Invention

The present invention relates to power amplifiers, and more particularlyto radio frequency (RF) power amplifiers employing high voltage and highpower metal-oxide semiconductor field-effect transistors (MOSFET).

2. Description of Related Art

High-power RF amplifiers adapted to operate over a range of 1-60 MHzwithout tuning have typically employed vacuum tubes. It would bedesirable to provide a high-power RF amplifier for operation over arange of 1-60 MHz, and which employs a minimum number of MOSFETtransistors instead of vacuum tubes.

BRIEF SUMMARY OF THE INVENTION

In accordance with one aspect of the invention, provided is an RF poweramplifier including a first field effect transistor having a first gate,a first source, and a first drain, having an output power rating of atleast 200 watts, and operating with a drain-to-source voltage that isgreater than 50 VDC. The amplifier includes a second field effecttransistor having a second gate, a second source, and a second drain,having an output power rating of at least 200 watts, and operating witha drain-to-source voltage that is greater than 50 VDC. The transistorsare configured as a push-pull amplifier. The amplifier further includesan RF signal input. An input transformer is connected to the RF signalinput. The input transformer has respective balanced outputs connectedto the first gate and the second gate. A broadband output transformerhas a first balanced input connected to the first drain, and a secondbalanced input connected to the second drain. The broadband outputtransformer has an input to output impedance ratio of 1:4 and at leastsome flux cancellation occurs within the broadband output transformer.Temperature compensating bias circuitry provides a temperaturecompensated bias voltage to the first field effect transistor and thesecond field effect transistor for decreasing the bias voltage of thefirst field effect transistor and the second field effect transistor astransistor temperature increases, in order to maintain a constant biascurrent level over temperature. The temperature compensating biascircuitry includes a temperature sensor generating a temperature signal.A first amplifier has an output providing a first temperature dependentvoltage based on the temperature signal. A second amplifier has anoutput providing a second temperature dependent voltage based on thetemperature signal. The first temperature dependent voltage and thesecond temperature dependent voltage change at substantially the samerate in response to the temperature signal. A potentiometer is connectedto the output of the first amplifier and the output of the secondamplifier such that a voltage across the potentiometer remainssubstantially constant when the first temperature dependent voltage andthe second temperature dependent voltage change. A bias output of thetemperature compensating bias circuitry is connected to at least one ofthe first field effect transistor and the second field effect transistorand supplies the temperature compensated bias voltage to the at leastone of the first field effect transistor and the second field effecttransistor.

In accordance with another aspect of the invention, provided is an RFpower amplifier including a first plurality of field effect transistorshaving directly interconnected drains and respective output powerratings of at least 100 watts, and a second plurality of field effecttransistors having directly interconnected drains and respective outputpower ratings of at least 100 watts. The transistors operate with adrain-to-source voltage that is greater than 50 VDC. The first pluralityof field effect transistors and the second plurality of field effecttransistors together form a push-pull amplifier having an output powerrating of at least 400 watts. The amplifier further includes an RFsignal input. An input transformer is connected to the RF signal input.The input transformer has respective balanced outputs connected to thegates of the transistors. A broadband output transformer has a firstbalanced input connected to the drains of the first plurality of fieldeffect transistors, and a second balanced input connected to the drainsof the second plurality of field effect transistors. The broadbandoutput transformer has an input to output impedance ratio of 1:4 and atleast some flux cancellation occurs within the broadband outputtransformer. Temperature compensating bias circuitry provides atemperature compensated bias voltage to the first plurality of fieldeffect transistors and the second plurality of field effect transistorsfor decreasing the bias voltage of the first plurality of field effecttransistors and the second plurality of field effect transistors astransistor temperature increases, in order to maintain a constant biascurrent level over temperature. The temperature compensating biascircuitry includes a temperature sensor generating a temperature signal.A first amplifier has an output providing a first temperature dependentvoltage based on the temperature signal. A second amplifier has anoutput providing a second temperature dependent voltage based on thetemperature signal. The first temperature dependent voltage and thesecond temperature dependent voltage change at substantially the samerate in response to the temperature signal. A potentiometer is connectedto the output of the first amplifier and the output of the secondamplifier such that a voltage across the potentiometer remainssubstantially constant when the first temperature dependent voltage andthe second temperature dependent voltage change. A bias output of thetemperature compensating bias circuitry is connected to at least one ofthe first plurality of field effect transistors and the second pluralityof field effect transistors and supplies the temperature compensatedbias voltage to the at least one of the first plurality of field effecttransistors and the second plurality of field effect transistors.

In accordance with another aspect of the invention, provided is an RFpower amplifier including a push-pull amplifier, the push-pull amplifiercomprising a first field effect transistor and a second field effecttransistor. Temperature compensating bias circuitry provides atemperature compensated bias voltage to the first field effecttransistor and the second field effect transistor for decreasing thebias voltage of the first field effect transistor and the second fieldeffect transistor as transistor temperature increases. The temperaturecompensating bias circuitry includes a temperature sensor generating atemperature signal. A first amplifier has an output providing a firsttemperature dependent voltage based on the temperature signal. Apotentiometer is connected to the output of the first amplifier suchthat a voltage across the potentiometer remains substantially constantwhen the first temperature dependent voltage changes. A bias output ofthe temperature compensating bias circuitry is connected to at least oneof the first field effect transistor and the second field effecttransistor and supplies the temperature compensated bias voltage to theat least one of the first field effect transistor and the second fieldeffect transistor.

In accordance with another aspect of the invention, provided is an RFpower amplifier including a push-pull amplifier, the push-pull amplifiercomprising a first field effect transistor having a first drain and asecond field effect transistor having a second drain. A broadband outputtransformer has a first balanced input and a second balanced input. Afirst DC-blocking transformer is connected between the first balancedinput and the first drain. A second DC-blocking transformer is connectedbetween the second balanced input and the second drain.

In accordance with another aspect, provided is an RF power amplifier.The RF power amplifier includes a first amplifier module and a secondamplifier module. The first amplifier module comprises a first push-pullamplifier including a plurality of field effect transistors and a firstoutput balun transformer. An output impedance of the first amplifiermodule is 25 ohms. The second amplifier module comprises a secondpush-pull amplifier including a plurality of field effect transistorsand a second output balun transformer. The output impedance of thesecond amplifier module is 25 ohms. A combiner is connected to the firstamplifier module and the second amplifier module. The combiner comprisesan unbalanced-to-unbalanced output transformer having an input-to-outputimpedance ratio of 1:4. The combiner combines an output from the firstamplifier module and an output from the second amplifier module into acombined signal. The combined signal is supplied to theunbalanced-to-unbalanced output transformer. An output impedance of thecombiner is 50 ohms.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a top-level schematic block diagram of a modular RF poweramplifier system;

FIG. 2 is a schematic circuit diagram of an RF power amplifier module;

FIG. 2 a is a schematic circuit diagram of a portion of FIG. 2;

FIG. 3 is a schematic circuit diagram of an RF power amplifier module;

FIG. 4 is a schematic circuit diagram of temperature compensating biascircuitry for an RF power amplifier module;

FIG. 5 is a schematic circuit diagram of temperature compensating biascircuitry for an RF power amplifier module;

FIG. 6 is a schematic circuit diagram of an RF power amplifier module;

FIG. 7 is a schematic diagram of DC-blocking transformers for an RFpower amplifier module;

FIG. 8 is a schematic circuit diagram of an RF power amplifier module;

FIG. 9 is a schematic circuit diagram of temperature compensating biascircuitry for an RF power amplifier module;

FIG. 10 is a schematic circuit diagram of an RF power amplifier module

FIG. 11 is a schematic circuit diagram of an RF power amplifier; and

FIG. 12 is a schematic circuit diagram of an RF power amplifier.

DETAILED DESCRIPTION OF THE INVENTION

As used herein, the terms “connected” and “connected to” refer aphysical and/or electrical joining or linking of one thing to another,and includes direct and indirect connections. For example, an amplifiercan be connected to an RF signal input by direct electrical connectionbetween the amplifier and input, or connected to said input via anindirect electrical connection, such as through an interposing resistoror capacitor. In the former case, the amplifier is directly connected tothe input. In the latter case, the amplifier is indirectly connected tothe input. However, in both cases, the amplifier is connected to the RFinput.

FIG. 1 shows a top level schematic block diagram of a modular RF poweramplifier system. The amplifier system includes an RF input terminal 10and an RF output terminal 11. In an embodiment, the characteristicimpedance of the input and output terminals is 50 ohms. The terminals10, 11 can be adapted for use with removable connectors. For example,the terminals 10, 11 can include a BNC or SMA female connector forremovably connecting to a BNC or SMA male connector, respectively.

An RF signal to be amplified by the amplifier system is provided at theRF input terminal 10. The RF signal is split by splitter 12 intoseparate signals to be amplified by a number of power amplifier modules13 a-13 d. In the example of FIG. 1, four power amplifier modules 13a-13 d are provided. Therefore, the splitter 12 splits the RF signalinto four separate signals. It is to be appreciated that fewer than fourpower amplifier modules could be provided, such as two or threeamplifier modules, for example. It is to be further appreciated thatgreater than four power amplifier modules could be provided, such as sixor eight amplifier modules, for example. The splitter 12 is designed tosplit the RF input signal into as many separate signals as there arepower amplifier modules. The splitter 12 can include a cascade ofseparate splitters for staged splitting of the RF input signal into anumber of separate signals.

The power amplifier modules 13 a-13 d amplify the separate signals fromthe splitter 12 according to the power ratings of the amplifier modules13 a-13 d. In an embodiment, each power amplifier module 13 a-13 d has apower rating of 400 watts. A modular RF power amplifier system employingfour 400 watt power amplifier modules would provide approximately 1600watts of total amplification. In another embodiment, each poweramplifier module 13 a-13 d has a power rating of 600 watts. A modular RFpower amplifier system employing four 600 watt power amplifiers wouldprovide approximately 2400 watts of total amplification. In stillanother embodiment, each power amplifier module 13 a-13 d has a powerrating of 1200 watts. A modular RF power amplifier system employing four1200 watt power amplifiers would provide approximately 4800 watts oftotal amplification. In still another embodiment, each power amplifiermodule 13 a-13 d has a power rating of 1500 watts. A modular RF poweramplifier system employing four 1500 watt power amplifies would provideapproximately 6000 watts of total amplification. As discussed above,fewer than four power amplifier modules could be provided or greaterthan four power amplifier modules could be provided. The number of poweramplifier modules to be used and their power ratings can be chosen basedon the required power output.

Example power amplifier modules 13 a-13 d are shown in FIGS. 2 and 3 andare discussed in detail below. In an embodiment, each power amplifiermodule 13 a-13 d is provided on a separate printed circuit board. Inother embodiments, a plurality of amplifier modules are provided on oneor more printed circuit boards.

Outputs from each of the power amplifier modules 13 a-13 d are providedto a combiner 14. In the example of FIG. 1, four power amplifier modules13 a-13 d are provided and the combiner 14 combines the four outputsfrom the amplifier modules into a single, combined RF output. Thecombiner 14 is designed to combine as many signals as there are poweramplifier modules. The combiner 14 can include a cascade of separatecombiners for staged combining of the amplified signals into a combinedsignal. As shown in FIG. 1, the RF output from the combiner 14 isprovided to the RF output terminal 11.

An RF power amplifier could comprise a single amplifier module, ratherthan a plurality of modules 13 a-13 d as shown in FIG. 1. An RF poweramplifier having a single amplifier module would not require thesplitter 12 and combiner 14.

FIG. 2 shows a push-pull power amplifier module 21. Transistors Q1 andQ2 form a push-pull pair for amplifying an RF signal.

The components that form the amplifier module 21 can be mounted on aprinted circuit board 22. The amplifier module includes an RF inputterminal 23 and an RF output terminal 24. In an embodiment, thecharacteristic impedance of the module's input and output terminals is50 ohms. The terminals 23, 24 can be adapted for use with removableconnectors, such as BNC or SMA connectors, for example.

An RF signal to be amplified by the amplifier module 21 is provided atthe RF input terminal 23. The RF signal is transmitted to an inputtransformer T1. The transformer T1 has an unbalanced or single-endedside, which is connected to the RF input terminal 23. The unbalanced orsingle-ended side is coupled to a balanced or differential side of thetransformer T1. The transformer T1 is shown as a so-called conventionaltransformer, having separated primary and secondary windings. It is tobe appreciated that the transformer T1 could be constructed as atransmission line transformer, which does not have separated primary andsecondary windings. An optional ground reference is provided for thetransformer's balanced or differential side via a resistor R1. Thetransformer T1 has an input to output impedance ratio of 4:1, forexample, and serves to divide the unbalanced RF signal into balancedsignals, 180° out of phase, for amplification by the push-pulltransistor pair Q1, Q2, and subsequent combination. It is to beappreciated that the transformer T1 could have an input to outputimpedance ratio of other than 4:1, such as 1:1.414, 1:9, 1:25, 3:2, etc.Further, it is to be appreciated that a so-called pi attenuator or piinput attenuator (not shown) comprising a plurality of resistors in aGreek letter “pi” configuration could be provided between the RF inputterminal 23 and the transformer T1, for normalizing gain of theamplifier module 21. Other attenuators could be provided between the RFinput terminal 23 and the transformer T1, such as an L attenuator or a Tattenuator, for example.

The input transformer T1 is shown as an unbalanced-to-balancedtransformer. However, it is to be appreciated that T1 couldalternatively be a balanced-to-balanced transformer.

An optional compensation capacitor C1 is connected across the balancedterminals of the transformer T1 and provides a low-pass filter responsewhich absorbs the inductance of T1 and helps compensate for a loss ofgain at higher frequencies.

The transistors Q1, Q2 that form the push-pull pair are high-voltageMOSFET RF power transistors. The transistors Q1, Q2 have output powerratings of at least 150 watts, preferably at least 200 watts, andoperate with a drain-to-source voltage that is greater than 50 VDC, suchas 62 VDC, 72 VDC, 86 VDC, 96 VDC or 100 VDC, for example. Exampletransistors Q1, Q2 have output power ratings of 150 watts, 300 watts,and 750 watts. An example 150 watt transistor is model ARF520manufactured by ADVANCED POWER TECHNOLOGY®. A further example 150 watttransistor is model SD3931 manufactured by STMICROELECTRONICS®. Anexample 300 watt transistor is model SD3933 manufactured bySTMICROELECTRONICS®. An example 750 watt transistor is model ARF1500manufactured by ADVANCED POWER TECHNOLOGY®. It is to be appreciated thattransistors having output power ratings other than 150 watts, 300 watts,and 750 watts can be used in an amplifier module as shown in FIG. 2. Forexample, 200 watt transistors can be used in the amplifier module.

An amplifier module 21 as shown in FIG. 2, having 150 watt transistorsQ1, Q2 forming a push-pull pair, can have a power rating of 300 watts.An amplifier module 21 having 300 watt transistors Q1, Q2 forming apush-pull pair, can have a power rating of 600 watts. An amplifiermodule 21 having 750 watt transistors Q1, Q2 forming a push-pull pair,can have a power rating of 1500 watts. It is to be appreciated that thepower rating of the amplifier module depends on the power rating of theselected transistors and the drain-to-source voltage at which thetransistors are operated. Further, it is to be appreciated thatamplifier modules of various power ratings can be constructed.

The balanced signals from the transformer T1 are respectively providedto the gates of the transistors Q1, Q2. One balanced signal is providedto the gate of transistor Q2 through a coupling capacitor C2 and aresistor R2, for amplification by the transistor Q2. The other balancedsignal is provided to the gate of transistor Q1 through a couplingcapacitor C3 and resistor R3, for amplification by the transistor Q1. Inan embodiment, capacitors C2 and C3 are each formed by two paralleled 47nF capacitors and resistors R2 and R3 are each formed by four paralleled15 ohm resistors.

A DC bias voltage for the gate of transistor Q2 is provided at BIAS2.The bias voltage is provided to the gate of transistor Q2 through an RFchoke RFC2 and resistor R4. A capacitor C4 is connected to the RF chokeRFC2 and resistor R4, and also to ground, and provides a low impedancepath to ground for high frequency signals. In an embodiment, the RFchoke RFC2 has a value of 10 μH. The DC bias voltage for the gate oftransistor Q2 can be provided by a temperature compensating biascircuit.

Similarly, a DC bias voltage for the gate of transistor Q1 is providedat BIAS1. The DC bias voltage BIAS1 is provided to the gate oftransistor Q1 through an RF choke RFC1 and resistor R5. A capacitor C5is connected to the RF choke RFC1 and resistor R5, and also to ground.The capacitor C5 provides a low impedance path to ground for highfrequency signals. In an embodiment, the RF choke RFC1 has a value of 10μH. The DC bias voltage for the gate of transistor Q1 can be provided bya temperature compensating bias circuit. Temperature compensating biascircuits are shown in FIG. 4, FIG. 5 and FIG. 9 and are discussedfurther below.

In an embodiment, capacitors C4 and C5 are each formed by two paralleled47 nF capacitors and resistors R4 and R5 are each formed by fourparalleled 110 ohm resistors.

The gates of the transistors Q1, Q2 are respectively connected to groundthrough resistors R7 and R6, which provide a discharge path for thecharge on the gates when DC bias is removed, and provide a solid groundreference for the DC bias voltages for the gates. The respective sourcesof transistors Q1 and Q2 are directly connected to ground. In anembodiment, resistors R6 and R7 each have a value of 10 kΩ.

As shown in FIG. 2, a DC power source that is greater than 50 VDC isconnected to the drain of each transistor Q1, Q2 through a common modechoke T2. Coils L1 and L2 of the common mode choke T2 are connected suchthat magnetic flux is cancelled during each RF cycle, to minimize thenet flux inside of the choke and, therefore, minimize the size of itscore. A coil L3, which is an additional winding of the common modechoke, provides a negative feedback signal from the DC feed structure.Coil L3 can be provided by a single turn through the center of the coreof the common mode choke T2. Negative feedback serves to lower the inputimpedance and to stabilize the amplifier's gain over its frequencyrange. A feedback path for the gate of transistor Q1 is provided througha network that includes resistor R9 and either capacitor C7 oroptionally capacitor C9. A feedback path for the gate of transistor Q2is provided through a network that includes resistor R10 and eithercapacitor C8 or optionally capacitor C10. In an embodiment, capacitorsC7 and C8 are each formed by two paralleled 47 nF capacitors whilecapacitors C9 and C10 are not used, and resistors R9 and R10 are eachformed by four paralleled 36 ohm resistors.

It is to be appreciated that coil L3 may be omitted and that negativefeedback may be taken directly from the drains of the transistors Q1 andQ2. In such a configuration, the feedback resistors R9 and R10 will bemade physically larger because the voltage at the drains isproportionally larger by the winding ratio of coil L3 to T2. Morespecifically, the voltage from coil L3 is proportional to the ratio ofL3 turns divided by the sum of the turns of coils L1 and L2 times the RFdrain-to-drain voltage applied to the primary low impedance side oftransformer T3 (transformer T3 is discussed in detail below). Forexample if coil L3 is one turn and L1 and L2 are each eight turns, thevoltage available from L3 is 1/16th of the drain-to-drain RF voltage.

The DC power source is connected to the common mode choke T2 through anetwork that includes inductors L4 and L5, a capacitor C6 and a resistorR8. The inductors L5 and L6 provide a high impedance to RF signals and ashort circuit for the DC power source. RF signals are decoupled from theDC power source by conduction to ground through the resistor R8 and thecapacitor C6. Inductors L4 and L5 can be ferrite bead inductors. In anembodiment, inductor L5 is a wound ferrite core with a value of 10 μH,and capacitor C6 is formed by six paralleled 47 nF capacitors. Inaddition to capacitor C6, similar grounded capacitors can be providedbetween inductor L4 and the DC power source, and between inductor L5 andthe common mode choke T2. If, in addition to capacitor C6, similargrounded capacitors are provided between inductor L5 and common modechoke T2, then resistor R8 can be omitted. Sometimes, in an effort toincrease isolation between the drains, the common DC feed point terminalof the common mode choke T2 is split, and each winding is fed byidentical decoupling networks similar to inductor L5, resistor R8,capacitor C6, and inductor L4. In such a configuration, the resistorsneed to be twice the value of resistor R8 because they are ACparalleled. If in addition to capacitor C6 similar grounded capacitorsare used at the DC feed terminals of the common mode choke T2, theresistors can be omitted.

A broadband transmission line output transformer T3 is connected to thedrain of each transistor Q1, Q2 and combines the amplified RF signalsfrom each transistor Q1, Q2. The transformer T3 is abalanced-to-balanced transformer having respective balanced inputs 25,26 connected to the drains of the transistors Q1, Q2. In FIG. 2,balanced input 25 is directly connected to the drain of transistor Q1,and balanced input 26 is directly connected to the drain of transistorQ2. However, it is to be appreciated that DC-blocking capacitors couldbe provided between the balanced inputs 25, 26 and the drains of thetransistors Q1, Q2. The transformer T3 has an input to output impedanceratio of 1:4 and performs an impedance matching function. In anembodiment, the characteristic impedance of the transistor outputcircuitry is 12.5 ohms. The transformer T3 combines the outputs from thetransistors Q1, Q2 and steps the characteristic impedance of the circuitup to 50 ohms.

Transformer T3 can be constructed using suitable cores, for example,toroid or ferrite tube cores, and coaxial cables having a characteristicimpedance of approximately or exactly 25 ohms. Each coaxial cable can bewound on its own core. 25 ohms is the geometric mean of a 12.5 ohm inputand a 50 ohm output. Performance of the transformer T3 is enhanced whenthe characteristic impedance of the transformer's cables is thegeometric mean of the input and output impedances. In an embodiment, thetransformer T3 includes coaxial cables having a characteristic impedanceof 25 ohms, a size 16 AWG stranded center conductor, TEFLON® insulationbetween an outer braid and the center conductor, and an insulatingjacket. The 25 ohm transmission line used to construct the transformerT3 can also be constructed by paralleling standard 50 ohm coaxial cable.Further, the transmission line used to construct the transformer T3 maybe constructed from parallel magnet wire or parallel twistedTEFLON®-insulated wire.

As stated above, the transformer T3 is a balanced-to-balancedtransformer. There is essentially zero net flux within the cores of thetransformer T3 because the currents in its transmission lines travel inopposite directions, which gives rise to identical fluxes of oppositesense. It is to be appreciated that in a balanced-to-unbalanced orunbalanced-to-unbalanced transformer, the flux cancellation is not ascomplete. Therefore, the cores of the transformer T3 can be made smallerthan those used in a balanced-to-unbalanced or unbalanced-to-unbalancedtransformer. Smaller transformer cores allow for shorter windings,thereby increasing the high frequency response of the transformer T3.Also, smaller transformer cores can accommodate a greater number orwindings, resulting in increased inductance, which extends the lowerfrequency range of the transformer, such as to 1 MHz for example. In anembodiment, transformer T3 has an operating frequency range of 1-60 MHz.

One benefit of using a 1:4 impedance ratio transmission line transformeras an RF output transformer, as shown in FIG. 2, is that such atransformer is less difficult to construct than other transformers. Atransformer having a 1:4 impedance ratio has an integer turns ratio of1:2 and, therefore, does not require a winding tap. Such a transformeris less difficult to construct than a transformer having an impedanceratio of 1:2, which has a non-integer turns ratio of 1:1.414. However,in order to utilize a 1:4 impedance ratio transformer as an RF outputtransformer in a 600 watt push-pull amplifier module, transistors thatoperate with a drain-to-source voltage that is greater than 50 VDC mustbe used, rather than conventional 50 VDC MOSFETs.

An optional compensation capacitor C11 is connected across the balancedinput terminals 25, 26 of the transformer T3, and provides a low-passfilter response by absorbing the inductance of T3 and any outputcapacitance of the transistors Q1, Q2. This helps compensate for gainslope reduction at the high end of the amplifier module's frequencyrange.

The balanced outputs of transformer T3 are connected to an optionaloutput balun transformer T4 through DC-blocking capacitors C12, C13.Capacitor C12 is connected between one output of transformer T3 and oneinput of balun transformer T4. Capacitor C13 is connected between theother output of transformer T3 and the other input of balun transformerT4. The capacitors C12, C13 and balun transformer T4 are connected inseries between transformer T3 and the RF output terminal 24. In theembodiment of FIG. 2, the capacitors C12, C13 are located at the output,50 ohms side of transformer T3, and between transformer T3 and baluntransformer T4. By locating the DC-blocking capacitors at the output, 50ohms side of transformer T3, rather than at the input side, thecapacitors C12, C13 can be designed to handle a lower current. Suchcapacitors may be less expensive than capacitors designed to handle ahigher current. The optional output balun transformer T4 has animpedance ratio of 1:1 and is connected to RF output terminal 24. In anembodiment, the output balun transformer T4 is constructed using 50 ohmcoaxial cable and a ferrite core. Alternatively, transformer T4 may beformed by paralleled magnet wire or by parallel or twistedTEFLON®-insulated wire. By locating the DC-blocking capacitors C12, C13between transformers T3 and T4, the RF output terminal 24 can bedirectly connected to the output balun transformer T4 and installeddirectly on a ground plane 27 of the printed circuit board 22. It is tobe appreciated that the output balun transformer T4 is optional and canbe omitted in some applications. However, omitting the output baluntransformer T4 may result in reduced efficiency and/or increasedintermodulation distortion (IMD).

A further schematic view of the broadband output transformer T3, theDC-blocking capacitors C12, C13, and the output balun transformer T4 isprovided at FIG. 2 a. In FIG. 2 a, the broadband output transformer T3and output balun transformer T4 are schematically shown as comprisingcoaxial cables. The output balun transformer T4 is shown having acoaxial cable shield connected to ground and a coaxial cable centerconductor connected to the RF output terminal 24. It is to beappreciated that the output balun transformer T4 can be reverselyconnected, so that the coaxial cable shield is connected to the RFoutput terminal 24 and the coaxial cable center conductor connected toground. The high impedance output side of the broadband outputtransformer T3 is connected the output balun transformer T4 through theDC-blocking capacitors C12, C13. In FIG. 2 a, at the high impedanceoutput side of transformer T3, the coaxial cable center conductors areshown as directly connected together, and the coaxial cable shields areconnected to respective DC-blocking capacitors C12, C13. It is to beappreciated that the high impedance output side of transformer T3 can bereversely connected, so that its coaxial cable shields are directlyconnected together, and the coaxial cable center conductors areconnected to respective DC-blocking capacitors C12, C13.

In an embodiment, capacitors C12 and C13 are each formed by sixparalleled 47 nF capacitors.

FIG. 3 shows another embodiment of a push-pull amplifier module 31.Various components shown in FIG. 3 are discussed above with respect toFIG. 2. Such components are referenced in FIG. 3 by identical referencecharacters as used in FIG. 2 and are not discussed in detail below.

A plurality of transistors having grounded sources and directlyinterconnected drains form each half of the push-pull amplifier. Forexample, two transistors Q3, Q4 form the “push” half of the amplifierand two transistors Q5, Q6 form the “pull” half of the amplifier. Thetransistors Q3-Q6 can have output power ratings of at least 100 wattsand operate with a drain-to-source voltage that is greater than 50 VDC.An example 100 watt transistor is model ARF463 manufactured by ADVANCEDPOWER TECHNOLOGY®. An amplifier module 31, having four 100 W transistorsQ3-Q6 can have a power rating of 400 W. An amplifier module 31 havingfour 150 watt transistors Q3-Q6 can have a power rating of 600 watts. Anamplifier module 31 having four 300 watt transistors Q3-Q6 can have apower rating of 1200 watts. In additional embodiments, each half of thepush-pull amplifier includes more than two transistors. It is to beappreciated that amplifier modules of various power ratings can beconstructed, based on the power rating of the selected transistors, thedrain-to-source voltage at which the transistors are operated, and thenumber of transistors provided in each half of the push-pull amplifier.

An RF signal to be amplified by the amplifier module 31 is provided atthe RF input terminal 23. The RF signal is transmitted to thetransformer T1 and split into balanced RF signals at the output oftransformer T1.

The balanced RF signals from transformer T1 are respectively provided totransformers T5 and T6. Transformers T5 and T6 are so-called Type 1Splitters, which further split the RF signals. The respective splitsignals from transformer T5 are provided to transformers T7 and T8,which are impedance matching transformers having an input-to-outputimpedance ratio of, for example, 1:4. The RF signal that is output fromimpedance matching transformer T7 is provided to the gate of transistorQ3 through DC-blocking capacitor C21 and resistor R21. Similarly, the RFsignal that is output from impedance matching transformer T8 is providedto the gate of transistor Q4 through DC-blocking capacitor C22 andresistor R22. In an embodiment, the DC-blocking capacitors C21 and C22are each formed by two paralleled 47 nF capacitors.

The respective split signals from transformer T6 are provided toimpedance matching transformers T9 and T10. The RF signal that is outputfrom impedance matching transformer T9 is provided to the gate oftransistor Q5 through DC-blocking capacitor C23 and resistor R23.Similarly, the RF signal that is output from impedance matchingtransformer T10 is provided to the gate of transistor Q6 throughDC-blocking capacitor C24 and resistor R24. In an embodiment, theDC-blocking capacitors C23 and C24 are each formed by two paralleled 47nF capacitors.

It is to be appreciated that a splitter utilizing resistors for furthersplitting the RF signals from transformer T1 could be used, rather thana Type 1 Splitter and impedance matching transformers as shown in FIG.3.

A DC bias voltage for the gate of transistor Q3 is provided at BIAS3 andthrough a resistor R25, for example, a 15Ω resistor. A capacitor C25,for example, a 47 nF capacitor, provides a low impedance path to groundfor high frequency signals. A DC bias voltage for the gate of transistorQ4 is provided at BIAS4 and through a resistor R26, for example, a 15Ωresistor. A capacitor C26, for example, a 47 nF capacitor, provides alow impedance path to ground for high frequency signals. A DC biasvoltage for the gate of transistor Q5 is provided at BIAS5 and through aresistor R27, for example, a 15Ω resistor. A capacitor C27, for example,a 47 nF capacitor, provides a low impedance path to ground for highfrequency signals. A DC bias voltage for the gate of transistor Q6 isprovided at BIAS6 and through a resistor R28, for example, a 15Ωresistor. A capacitor C28, for example, a 47 nF capacitor, provides alow impedance path to ground for high frequency signals.

Feedback signals are provided from the common mode choke T2 via coil L3through resistors R27 and R28. It is to be appreciated that negativefeedback could be taken directly from the drains of the transistorsQ3-Q6, which would require physically larger dissipation resistors R27,R28. In an embodiment, resistors R27 and R28 are each formed by twoparalleled 430 ohm resistors.

Resistors R29-R34 are balancing resistors for absorbing amplitudeimbalance between the signal splitter outputs due to productiontolerances. The resistors R29-R34 also help to maintain correct inputport impedance, contributing to a low voltage standing wave ratio(VSWR). In an embodiment, resistors R29 and R32 are each formed by twoparalleled 51 ohm resistors, and resistors R30, R31, R33, and R34 areeach formed by two paralleled 15 ohm resistors.

Resistors R35-R38 provide discharge paths for the charge on the gateswhen DC bias is removed, and provide solid ground references for the DCbias voltages for the gates. In an embodiment, resistors R35-R38 eachhave a value of 10 kΩ.

The drains of transistors Q3 and Q4 are interconnected or directlyconnected together and are directly connected to an input 26 oftransformer T3. The drains of transistors Q5 and Q6 are alsointerconnected or directly connected together and are directly connectedto an input 25 of transformer T3. The “push” signal from transistors Q3and Q4 are combined with the “pull” signal from transistors Q5 and Q6 bytransformer T3.

FIG. 4 shows temperature compensating bias circuitry 40 for DC biasingthe transistors that form the push-pull amplifiers discussed herein. Inthe amplifier shown in FIG. 2, the DC bias voltage from the temperaturecompensating bias circuitry 40 would be provided at BIAS1 and BIAS2. Inthe amplifier shown in FIG. 3, the DC bias voltage from the temperaturecompensating bias circuitry 40 would be provided at BIAS3 through BIAS6.

The temperature of the transistors in the push-pull amplifiers willincrease during operation of the amplifiers. To maintain a lineartransistor operation at a constant idle point (bias current level), thetemperature compensating bias circuitry 40 reduces the bias voltage ofthe transistors as their temperature increases. In an embodiment, thetemperature compensating bias circuitry 40 is designed to reduce thebias voltage at a rate of 6.35 mV/° C.

The temperature compensating bias circuitry 40 includes a temperaturesensor Q41. The output of the temperature sensor Q41 changes withtemperature. The temperature sensor Q41 can be mounted so as to sensethe temperature of the transistors in the push-pull amplifier. Forexample, the temperature sensor Q41 can be attached to a heat sink forthe transistors. In FIG. 4, the temperature sensor Q41 is a transistorand the temperature sensor provides a decreasing temperature signallevel (e.g., −2.1 mV/° C.) as temperature increases. Alternatively, thetemperature sensor could provide an increasing temperature signal level(e.g., 10 mV/° C.) as temperature increases. An example temperaturesensor is a TIP31C power transistor. Another example temperature sensoris a temperature sensor integrated circuit (IC), such as an LM35 sensorfrom NATIONAL SEMICONDUCTOR®. It is to be appreciated that other typesof temperature sensors could be incorporated into the temperaturecompensating bias circuitry 40, such as resistive temperature devices(RTDs) or thermistors for example, and that the temperature compensatingbias circuitry can include a plurality of temperature sensors mounted atdifferent locations on or around the push-pull amplifier.

The temperature compensating bias circuitry 40 further includes a firstamplifier U41, a second amplifier U42, a plurality of resistors R41-R51,and a potentiometer R52. The first and second amplifiers U41, U42 can beoperational amplifiers, such as the LM324A op-amp. The wiper of thepotentiometer R52 provides a bias output to supply the temperaturecompensated DC bias voltage to the transistors of the push-pullamplifier (not shown in FIG. 4). In an embodiment, the potentiometer R52is a 5 kΩ potentiometer. Multiple potentiometers can be paralleled toprovide individual bias control to each transistor in the push-pullamplifier. The first amplifier U41 is configured to provide a firsttemperature dependent voltage to the potentiometer R52 based on thetemperature signal from the temperature sensor Q41. The second amplifierU42 is configured to provide a second temperature dependent voltagebased on the temperature signal to the potentiometer R52 throughresistor R46. The first temperature dependent voltage and the secondtemperature dependent voltage change at substantially the same rate inresponse to the temperature signal. For example, both of the firsttemperature dependent voltage and the second temperature dependentvoltage decrease at a rate of 6.35 mV/° C. Although the outputs of theamplifiers U41, U42 change with increasing temperature, the voltageacross the potentiometer remains substantially constant, which allowsthe full 6.35 mV/° C. voltage reduction to be provided, at any physicalsetting of the potentiometer, as reduced bias voltage to the push-pullamplifier's transistors (rather than a voltage divided percentage of6.35 mV/° C. as established by the wiper setting of the potentiometerR52). An example first temperature dependent voltage from the firstamplifier U41 is 5.0 volts at 25° C. minus 6.35 mV/° C. An examplesecond temperature dependent voltage from the second amplifier U42 is2.0 volts at 25° C. minus 6.35 mV/° C.

An example value for resistors R41, R42, R45 and R48 is 10.0 kΩ. Anexample value for resistor R43 is 16.6 kΩ. An example value for resistorR44 is 24.9 kΩ. An example value for resistor R46 is 1 kΩ. An examplevalue for resistor R47 is 430 kΩ. An example value for resistor R49 is4.99 kΩ. An example value for resistor R50 is 2 kΩ. An example value forresistor R51 is 3.3 kΩ. A 5V input is shown as provided to resistorsR43, R47 and R50. The 5V input can be a programmable reference voltageand is not limited to 5V. In an embodiment, the 5V input is provided byan adjustable precision voltage reference, such as an LM385 availablefrom NATIONAL SEMICONDUCTOR®.

FIG. 5 shows a further example of temperature compensating biascircuitry 50 for DC biasing the transistors that form the push-pullamplifiers discussed herein. The bias circuitry 50 includes anadjustable precision voltage reference D51 and a temperature sensor Q51.With a 12 VDC supply 56 as shown, the bias circuitry is configured tosupply a substantially constant 2.5 VDC across potentiometers R52 a andR52 b, while the first temperature dependent voltage 53 is 5.0 volts at25° C. minus 6.35 mV/° C. and the second temperature voltage 54 is 2.0volts at 25° C. minus 6.35 mV/° C. Supply 56 voltages other than 12 VDC(e.g., 8 VDC) could be used by the bias circuitry 50. The bias circuitry50 includes two bias outputs BIAS1, BIAS2 for supplying the temperaturecompensated DC bias voltage to the transistors of the push-pullamplifier (not shown in FIG. 5). Example component values for resistorsand capacitors in the bias circuitry 50 are shown in the figure. In FIG.5, amplifiers U41 and U42 feed transistors Q53 and Q54, respectively,rather than feeding the potentiometers R52 a and R52 b directly.Transistors Q53 and Q54 are configured as followers and allow theamplifiers U41 and U42 to supply the necessary current when multiplepotentiometers are used. An optional diode D52 is provided at the outputof amplifier U42 to compensate for any ambient temperature-inducedchanges in the operation of the transistors Q53 and Q54. Multiple diodescan be used instead of a single diode D52 as shown, to compensate forambient temperature-induced changes in the operation of othercomponents, such as changes in the operation of the diodes locatedbetween the potentiometers R52 a, R52 b and the bias outputs BIAS1,BIAS2.

The temperature compensating bias circuitry 50 of FIG. 5 includes anamplifier U51 that generates a temperature output signal 55. Thetemperature output signal 55 transmits the temperature of the RF poweramplifier module and can be monitored by a remote device and/or used todisplay a current temperature of the RF power amplifier module. The biascircuitry 50 can be configured so that the temperature output signal 55increases at a constant rate, such as 50 mV/° C., or decreases at aconstant rate over a voltage range (e.g., 0-5 VDC) corresponding to atemperature range (e.g., 2° C.-102° C.). The bias circuitry 50 canfurther be configured so that the temperature output signal 55 increasesor decreases at a constant rate per bit of an analog to digital (A/D)converter (e.g., 1° F. per bit for an 8 bit ND). In FIG. 5, thetemperature output signal 55 is configured to be 5 VDC at 11.5° F. and 0VDC at 266.5° F., which is 1° F. per bit for an 8 bit ND. It is to beappreciated that the amplifier U51 can be configured to generate apositive going signal suitable for display on an analog meter.

The temperature compensating bias circuitry 50 of FIG. 5 furtherincludes an amplifier U52 (e.g., an operational amplifier) configured asa comparator that controls activations of transistor Q52 based on thetemperature signal from the temperature sensor Q51. When the temperatureexceeds a set point or limit established by the voltage dividerconsisting of the 18K and 11.5K resistors, the comparator output causesthe transistor Q52 to conduct. This generates a high temperature alarmon a T LIMIT line 57, to signal an off board controller (not shown) thata fault condition exists and that corrective action needs to be taken. ATRANSISTOR DUMP line 58 allows the off board controller to deactivatethe transistors of the push-pull amplifier. The off board controller candeactivate the transistors of the push-pull amplifier based on monitoredparameters, such as the temperature of the amplifier module. In theevent of any fault, it is desirable for the RF transistors to stopamplifying as soon as possible. One way to do this is to shut thetransistors off by removing bias and discharging the gates as soon aspossible. Activation of a dump transistor Q55 pulls the gates of thepush-pull amplifier to ground through 10Ω resistors, which limit thedischarge current to a safe level that will not damage transistor Q55.

It is to be appreciated that the use of quad op-amp ICs in thetemperature compensating bias circuitry 50 will provide amplifiers U41,U42, U51 and U52 in a single package. The marginal cost increase of aquad op-amp IC versus a dual op-amp IC is typically small. Therefore,although amplifiers U51 and U52 are optional elements of the temperaturecompensating bias circuitry 50, their addition should not increase thecost of the circuitry 50 very much, rendering the temperature output andthe temperature alarm signals inexpensive to generate.

FIG. 9 shows a further example of temperature compensating biascircuitry 90 for DC biasing the transistors that form the push-pullamplifiers discussed herein. The bias circuitry 90 in FIG. 9 includes atemperature sensor IC 91 rather than using a transistor as a temperaturesensor. The temperature sensor IC 91 provides an increasing temperaturesignal level as the temperature of the RF module increases (e.g., 10mV/° C.). The bias circuitry 90 processes the increasing temperaturesignal level from the temperature sensor IC 91 to provide the DC biasoutputs BIAS1 and BIAS2 for the transistors in the push-pull amplifiers(not shown), the temperature output signal 55, and the high temperaturealarm 57. As compared to the bias circuitry 50 shown in FIG. 5,transistors Q53 and Q54 are omitted from the bias circuitry 90 shown inFIG. 9. However, the wipers of the potentiometers R52 a, R52 b areconnected to transistors Q92 and Q93, which function as followers andgenerate the DC bias outputs BIAS1, BIAS2. The emitters of the followertransistors Q92, Q93 are grounded through a low value resistor toprovide a low impedance source characteristic that provides low IMD.Further, in the bias circuitry 90 of FIG. 9, the dump transistor Q55 isconnected to the “+” terminal of amplifier U41. When the dump transistorQ55 is activated by the TRANSISTOR DUMP line 58, it essentially forcesamplifier U41 to reduce its output to 0V, causing the voltage at thebias outputs to drop to a very low value below the gate thresholdvoltage of the push-pull amplifier transistors. The gates of the RFMOSFETs in the push-pull amplifiers then discharge through gateresistors (e.g., see resistors R6 and R7 in FIG. 2). Because thetransistor followers allow a very large amount of current to be appliedto the RF transistor gate resistors, the gate resistors can be changedto a much lower value (example from 10 KΩ to 470Ω or even 100Ω) todecrease the discharge time of the RF transistor gates. The addedbenefit of the lower value gate resistor is an improvement in the IMD ofthe amplifier module.

FIG. 6 and FIG. 8 show modifications of the push-pull power amplifiermodule 21 of FIG. 2. Various components shown in FIG. 6 and FIG. 8 arediscussed above with respect to FIG. 2. Such components are referencedin FIG. 6 and FIG. 8 by identical reference characters as used in FIG. 2and are not discussed in detail below.

In FIG. 6, the DC-blocking capacitors C12, C13 are replaced withDC-blocking transformers T61 and T62, and the common mode choke T2 isomitted. The DC-blocking capacitors C12, C13 can be a source of failurein the power amplifier module 21 when they are subject to very high RFcurrents. Performance and reliability of the power amplifier module 21can be improved by using DC-blocking transformers rather thancapacitors.

The DC-blocking transformers T61, T62 are respectively connected betweenthe drains of transistors Q1 and Q2 and the balanced inputs 25, 26 ofthe broadband output transformer T3. A negative feedback signal from theDC feed structure is provided by the DC-blocking transformers T61, T62as shown. The negative feedback can be provided by a single turn of wirethrough the centers of the cores of the DC-blocking transformers T61,T62. See FIG. 7, which provides a schematic diagram of the DC-blockingtransformers T61, T62.

Optional matching coils LM1 and LM2 can be located between theDC-blocking transformers T61, T62 and the drains of the transistors Q1,Q2, and optional capacitors C61 and C62 can be located across therespective inputs of the DC-blocking transformers T61, T62. A bypasscapacitor C63 can be located at the DC power source. Optional capacitorC6 provides further bypassing and can help improve balance of thesystem.

FIG. 7 schematically shows the construction of the DC-blockingtransformers T61, T62. The DC-blocking transformers can be formed bywinding coaxial cables 71, 72 on respective cores 73, 74. In anembodiment, the coaxial cables have a characteristic impedance of 25ohms, which is twice the drain-drain differential impedance of thetransistors Q1, Q2. Connections to the center and shield conductors ofthe coaxial cables 71, 72 are shown in FIG. 7. The center and shieldconductors are further identified in FIG. 6 by “C” and “S”,respectively. The negative feedback loop 75 is also shown in FIG. 7. Itis to be appreciated that the center and shield connections can betransposed, and that the construction is not limited to coaxial cable(e.g., twisted wire pairs, paralleled wire, etc. can also be used). Itis to be further appreciated that power and current handling capabilityof said 25Ω cable can be greatly increased by constructing the 25Ω cableby paralleling two 50Ω cables, preferably connecting the shield of oneto the center of the other.

The input side of the DC-blocking transformers T61, T62 is identified as“Balanced RF In” and the output side is identified as “Balanced RF Out.”The shields of the coaxial cables 71, 72 are connected to ground at theinput side and, at the output side, provide the Balanced RF Out. Thecenter conductors are connected to the drains of the transistors Q1, Q2through optional matching coils LM1, LM2 (see FIG. 6). The optionalcapacitors C61, C62 (see FIG. 6) are each connected between a centerconductor and shield at the input side of the DC-blocking transformersT61, T62. The DC power source for the transistors Q1, Q2 is connected tothe center conductors of the cables 71, 72 at the output side of thetransformers T61, T62. The shields of the cables 71, 72 provide the RFoutput signal to the broadband output transformer T3 (see FIG. 6) at theoutput side of the transformers T61, T62.

It is to be appreciated that the DC-blocking transformers T61, T62 canbe constructed from paralleled or twisted wire if desired.

FIG. 8 shows the push-pull power amplifier module 21 of FIG. 6 with theaddition of transformer T2 a. Transformer T2 a generates the negativefeedback signal for the transistors Q1, Q2, instead of the DC-blockingtransformers T61, T62. Transformer T2 a can be connected on either sideof the optional matching coils LM1 and LM2. In an embodiment, theprimary of transformer T2 a is a single turn loop. Optional capacitorsC81 and C82 are also shown connected on either side of the output baluntransformer T4.

In a configuration differing from FIG. 6, FIG. 10 shows the DC-blockingtransformers connected to a 12.5 ohm characteristic impedance, 1:1impedance ratio balun T101, followed by a 1:4 impedance ratiounbalanced-to-unbalanced transformer T102. The balun T101 is a broadbandoutput transformer. Transformers T3 and T4 are eliminated in FIG. 10.

As discussed above, the amplifier modules described herein can becombined to create higher-powered amplifiers. It is to be appreciatedthat the combiners used to combine the outputs of the amplifier modulescan have a variety of output impedances, such as 12.5 ohm, 25 ohm, 50ohm, 100 ohm, etc. The output impedance of the combiners can further bematched to a desired characteristic impedance, such as 50 ohm, using atransformer that provides the proper impedance matching.

FIG. 11 and FIG. 12 both show two amplifier modules being combined tocreate a higher-powered amplifier. A splitter 103 splits an input signalto two push-pull amplifier modules. The amplifier modules in FIG. 11include DC-blocking capacitors C12, C13, whereas the amplifier modulesin FIG. 12 include DC-blocking transformers T61, T62. In FIG. 11 andFIG. 12, the output of each push-pull transistor pair in a module is fedto an output balun transformer T104, T105. The output balun transformersT104, T105 can have an input-to-output impedance ratio of 1:1. Theoutput balun transformers provide an output impedance of 25 ohms foreach amplifier module. The output balun transformers T104, T105 in FIG.11 supply a combiner 106. The combiner 106 combines the outputs from theamplifier modules into a combined signal. The combiner 106 includes atype 2 hybrid 0° 2-way combiner 107, which supplies anunbalanced-to-unbalanced output transformer 108 having aninput-to-output impedance ratio of 1:4. The output impedance of the type2 hybrid 0° 2-way combiner 107 is 12.5 ohms, which is stepped up to a 50ohm output impedance by the unbalanced-to-unbalanced output transformer108. The type 2 hybrid 0° 2-way combiner 107 supplies the combinedsignal to the unbalanced-to-unbalanced output transformer 108.

The combiner 109 in FIG. 12 is similar to the combiner in FIG. 11,except that the combiner 109 includes a type 1 0° hybrid combiner.

Example applications with which the disclosed modular RF power amplifiersystem may be used include radio communications, such as amateur radiocommunications, military radio communications, marine radiocommunications (e.g., ship to shore), high frequency radio telephonecommunications, and short wave radio broadcast stations.

It should be evident that this disclosure is by way of example and thatvarious changes may be made by adding, modifying or eliminating detailswithout departing from the fair scope of the teaching contained in thisdisclosure. The invention is therefore not limited to particular detailsof this disclosure except to the extent that the following claims arenecessarily so limited.

1. An RF power amplifier, comprising: a first field effect transistor:having a first gate, a first source, and a first drain, having an outputpower rating of at least 200 watts, and operating with a drain-to-sourcevoltage that is greater than 50 VDC; a second field effect transistor:having a second gate, a second source, and a second drain, having anoutput power rating of at least 200 watts, and operating with adrain-to-source voltage that is greater than 50 VDC; wherein saidtransistors are configured as a push-pull amplifier; an RF signal input;an input transformer connected to the RF signal input, the inputtransformer having respective balanced outputs connected to the firstgate and the second gate; a broadband output transformer having a firstbalanced input connected to the first drain, and a second balanced inputconnected to the second drain, wherein the broadband output transformerhas an input to output impedance ratio of 1:4, and further wherein atleast some flux cancellation occurs within the broadband outputtransformer; and temperature compensating bias circuitry for providing atemperature compensated bias voltage to the first field effecttransistor and the second field effect transistor for decreasing thebias voltage of the first field effect transistor and the second fieldeffect transistor as transistor temperature increases, the temperaturecompensating bias circuitry comprising: a temperature sensor generatinga temperature signal; a first amplifier having an output providing afirst temperature dependent voltage based on the temperature signal; asecond amplifier having an output providing a second temperaturedependent voltage based on the temperature signal, wherein the firsttemperature dependent voltage and the second temperature dependentvoltage change at substantially the same rate in response to thetemperature signal; and a potentiometer connected to the output of thefirst amplifier and the output of the second amplifier such that avoltage across the potentiometer remains substantially constant when thefirst temperature dependent voltage and the second temperature dependentvoltage change; and a bias output connected to at least one of the firstfield effect transistor and the second field effect transistor andsupplying the temperature compensated bias voltage to the at least oneof the first field effect transistor and the second field effecttransistor.
 2. A modular RF power amplifier system, comprising: a firstamplifier module comprising the RF power amplifier as set forth in claim1; and a second amplifier module comprising the RF power amplifier asset forth in claim 1; wherein outputs from said modules are combined toform a combined RF output.
 3. The RF power amplifier of claim 1, furthercomprising: an RF output terminal; an output balun transformer; and aDC-blocking capacitor, wherein the output balun transformer and theDC-blocking capacitor are connected in series between the broadbandoutput transformer and the RF output terminal, wherein the broadbandoutput transformer is a transmission line transformer including a cablehaving a characteristic impedance of approximately or exactly 25 ohms,wherein the broadband output transformer has balanced outputs and duringoperation of the RF power amplifier a net flux within the broadbandoutput transformer is substantially zero, and wherein the output baluntransformer has an input to output impedance ratio of 1:1 and acharacteristic impedance of approximately or exactly 50 ohms.
 4. An RFpower amplifier, comprising: a first plurality of field effecttransistors having directly interconnected drains and respective outputpower ratings of at least 100 watts; a second plurality of field effecttransistors having directly interconnected drains and respective outputpower ratings of at least 100 watts; wherein said transistors operatewith a drain-to-source voltage that is greater than 50 VDC, and whereinthe first plurality of field effect transistors and the second pluralityof field effect transistors together form a push-pull amplifier havingan output power rating of at least 400 watts; an RF signal input; aninput transformer connected to the RF signal input, the inputtransformer having respective balanced outputs connected to gates of thefirst plurality of field effect transistors and gates of the secondplurality of field effect transistors; and a broadband outputtransformer having a first balanced input connected to the drains of thefirst plurality of field effect transistors, and a second balanced inputconnected to the drains of the second plurality of field effecttransistors, wherein the broadband output transformer has an input tooutput impedance ratio of 1:4, and further wherein at least some fluxcancellation occurs within the broadband output transformer; andtemperature compensating bias circuitry for providing a temperaturecompensated bias voltage to the first plurality of field effecttransistors and the second plurality of field effect transistors fordecreasing the bias voltage of the first plurality of field effecttransistors and the second plurality of field effect transistors astransistor temperature increases, the temperature compensating biascircuitry comprising: a temperature sensor generating a temperaturesignal; a first amplifier having an output providing a first temperaturedependent voltage based on the temperature signal; a second amplifierhaving an output providing a second temperature dependent voltage basedon the temperature signal, wherein the first temperature dependentvoltage and the second temperature dependent voltage change atsubstantially the same rate in response to the temperature signal; and apotentiometer connected to the output of the first amplifier and theoutput of the second amplifier such that a voltage across thepotentiometer remains substantially constant when the first temperaturedependent voltage and the second temperature dependent voltage change;and a bias output connected to at least one of the first plurality offield effect transistors and the second plurality of field effecttransistors and supplying the temperature compensated bias voltage tothe at least one of the first plurality of field effect transistors andthe second plurality of field effect transistors.
 5. A modular RF poweramplifier system, comprising: a first amplifier module comprising the RFpower amplifier as set forth in claim 4; and a second amplifier modulecomprising the RF power amplifier as set forth in claim 4, whereinoutputs from said modules are combined to form a combined RF output. 6.The RF power amplifier of claim 4, further comprising: an RF outputterminal; an output balun transformer; and a DC-blocking capacitor,wherein the output balun transformer and the DC-blocking capacitor areconnected in series between the broadband output transformer and the RFoutput terminal, wherein the broadband output transformer is atransmission line transformer including a cable having a characteristicimpedance of approximately or exactly 25 ohms, wherein the broadbandoutput transformer has balanced outputs and during operation of the RFpower amplifier a net flux within the broadband output transformer issubstantially zero, and wherein the output balun transformer has aninput to output impedance ratio of 1:1 and a characteristic impedance ofapproximately or exactly 50 ohms.
 7. An RF power amplifier, comprising:a push-pull amplifier, the push-pull amplifier comprising a first fieldeffect transistor and a second field effect transistor; and temperaturecompensating bias circuitry for providing a temperature compensated biasvoltage to the first field effect transistor and the second field effecttransistor for decreasing the bias voltage of the first field effecttransistor and the second field effect transistor as transistortemperature increases, the temperature compensating bias circuitrycomprising: a temperature sensor generating a temperature signal; afirst amplifier having an output providing a first temperature dependentvoltage based on the temperature signal; and a potentiometer connectedto the output of the first amplifier such that a voltage across thepotentiometer remains substantially constant when the first temperaturedependent voltage changes; and a bias output connected to at least oneof the first field effect transistor and the second field effecttransistor and supplying the temperature compensated bias voltage to theat least one of the first field effect transistor and the second fieldeffect transistor.
 8. The RF power amplifier of claim 7, wherein thetemperature compensating bias circuitry further comprises a secondamplifier having an output providing a second temperature dependentvoltage based on the temperature signal, wherein the first temperaturedependent voltage and the second temperature dependent voltage change atsubstantially the same rate in response to the temperature signal, andwherein the potentiometer is connected to the output of the firstamplifier and the output of the second amplifier such that a voltageacross the potentiometer remains substantially constant when the firsttemperature dependent voltage and the second temperature dependentvoltage change.
 9. The RF power amplifier of claim 8, further comprisinga follower transistor connected to the potentiometer, wherein thefollower transistor generates the bias output.
 10. The RF poweramplifier of claim 7, wherein the temperature sensor is a transistor.11. The RF power amplifier of claim 7, wherein the temperature sensorcomprises a temperature sensor integrated circuit.
 12. The RF poweramplifier of claim 7, further comprising a temperature output signaltransmitting a temperature of the RF power amplifier.
 13. The RF poweramplifier of claim 7, further comprising a transistor dump signal fordeactivating at least one of the first field effect transistor and thesecond field effect transistor.
 14. The RF power amplifier of claim 7,wherein the first field effect transistor includes a first gate, a firstsource, and a first drain, wherein the second field effect transistorincludes a second gate, a second source, and a second drain, the RFpower amplifier further comprising: a broadband output transformerhaving a first balanced input and a second balanced input, wherein thebroadband output transformer has an input to output impedance ratio of1:4, and further wherein at least some flux cancellation occurs withinthe broadband output transformer; a first DC-blocking transformerconnected between the first balanced input and the first drain; and asecond DC-blocking transformer connected between the second balancedinput and the second drain.
 15. The RF power amplifier of claim 14,wherein each one of the broadband output transformer, the firstDC-blocking transformer, and the second DC-blocking transformer includea cable having a characteristic impedance of approximately or exactly 25ohms.
 16. The RF power amplifier of claim 14, wherein the firstDC-blocking transformer provides a negative feedback signal to the firstfield effect transistor, and wherein the second DC-blocking transformerprovides a negative feedback signal to the second field effecttransistor.
 17. The RF power amplifier of claim 14, further comprisinganother transformer, which provides negative feedback signals to thefirst field effect transistor and the second field effect transistor.18. The RF power amplifier of claim 14, further comprising an RF outputterminal; and an output balun transformer connected between thebroadband output transformer and the RF output terminal.
 19. An RF poweramplifier, comprising: a push-pull amplifier, the push-pull amplifiercomprising a first field effect transistor having a first drain and asecond field effect transistor having a second drain; a broadband outputtransformer having a first balanced input and a second balanced input; afirst DC-blocking transformer connected between the first balanced inputand the first drain; and a second DC-blocking transformer connectedbetween the second balanced input and the second drain.
 20. The RF poweramplifier of claim 19, further comprising temperature compensating biascircuitry for providing a temperature compensated bias voltage to thefirst field effect transistor and the second field effect transistor fordecreasing the bias voltage of the first field effect transistor and thesecond field effect transistor as transistor temperature increases. 21.The RF power amplifier of claim 19, wherein each one of the broadbandoutput transformer, the first DC-blocking transformer, and the secondDC-blocking transformer include a cable having a characteristicimpedance of approximately or exactly 25 ohms.
 22. The RF poweramplifier of claim 19, wherein the first DC-blocking transformerprovides a negative feedback signal to the first field effecttransistor, and wherein the second DC-blocking transformer provides anegative feedback signal to the second field effect transistor.
 23. TheRF power amplifier of claim 19, further comprising another transformer,which provides negative feedback signals to the first field effecttransistor and the second field effect transistor.
 24. The RF poweramplifier of claim 19, wherein the broadband output transformer has aninput to output impedance ratio of 1:4, and further wherein at leastsome flux cancellation occurs within the broadband output transformer.25. The RF power amplifier of claim 24, further comprising an RF outputterminal; and an output balun transformer connected between thebroadband output transformer and the RF output terminal.
 26. The RFpower amplifier of claim 19, wherein the broadband output transformerhas unbalanced outputs and an input to output impedance ratio of 1:1.27. The RF power amplifier of claim 26, further comprising an RF outputterminal; and an unbalanced-to-unbalanced transformer connected betweenthe broadband output transformer and the RF output terminal.
 28. The RFpower amplifier of claim 27, wherein the unbalanced-to-unbalancedtransformer has an input to output impedance ratio of 1:4.
 29. An RFpower amplifier, comprising: a first amplifier module comprising a firstpush-pull amplifier including a plurality of field effect transistorsand a first output balun transformer, wherein an output impedance of thefirst amplifier module is 25 ohms; a second amplifier module comprisinga second push-pull amplifier including a plurality of field effecttransistors and a second output balun transformer, wherein an outputimpedance of the second amplifier module is 25 ohms; a combinerconnected to the first amplifier module and the second amplifier module,the combiner comprising an unbalanced-to-unbalanced output transformerhaving an input-to-output impedance ratio of 1:4, wherein the combinercombines an output from the first amplifier module and an output fromthe second amplifier module into a combined signal, wherein the combinedsignal is supplied to the unbalanced-to-unbalanced output transformer,and wherein an output impedance of the combiner is 50 ohms.
 30. The RFpower amplifier of claim 29, wherein the first amplifier module includesa plurality of DC-blocking transformers, and wherein the secondamplifier module includes a plurality of DC-blocking transformers.